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Low Noise, 90 MHz Variable Gain Amplifier AD603
FEATURES
Linear-in-dB gain control Pin programmable gain ranges -11 dB to +31 dB with 90 MHz bandwidth 9 dB to 51 dB with 9 MHz bandwidth Any intermediate range, for example -1 dB to +41 dB with 30 MHz bandwidth Bandwidth independent of variable gain 1.3 nV/Hz input noise spectral density 0.5 dB typical gain accuracy
impedance (50 M), low bias (200 nA) differential input; the scaling is 25 mV/dB, requiring a gain control voltage of only 1 V to span the central 40 dB of the gain range. An overrange and underrange of 1 dB is provided whatever the selected range. The gain control response time is less than 1 s for a 40 dB change. The differential gain control interface allows the use of either differential or single-ended positive or negative control voltages. Several of these amplifiers may be cascaded and their gain control gains offset to optimize the system S/N ratio. The AD603 can drive a load impedance as low as 100 with low distortion. For a 500 load in shunt with 5 pF, the total harmonic distortion for a 1 V sinusoidal output at 10 MHz is typically -60 dBc. The peak specified output is 2.5 V minimum into a 500 load. The AD603 uses a patented proprietary circuit topology--the X-AMP(R). The X-AMP comprises a variable attenuator of 0 dB to -42.14 dB followed by a fixed-gain amplifier. Because of the attenuator, the amplifier never has to cope with large inputs and can use negative feedback to define its (fixed) gain and dynamic performance. The attenuator has an input resistance of 100 , laser trimmed to 3%, and comprises a seven-stage R-2R ladder network, resulting in an attenuation between tap points of 6.021 dB. A proprietary interpolation technique provides a continuous gain control function which is linear in dB. The AD603 is specified for operation from -40C to +85C.
APPLICATIONS
RF/IF AGC amplifier Video gain control A/D range extension Signal measurement
GENERAL DESCRIPTION
The AD603 is a low noise, voltage-controlled amplifier for use in RF and IF AGC systems. It provides accurate, pin selectable gains of -11 dB to +31 dB with a bandwidth of 90 MHz or 9 dB to 51 dB with a bandwidth of 9 MHz. Any intermediate gain range may be arranged using one external resistor. The input referred noise spectral density is only 1.3 nV/Hz and power consumption is 125 mW at the recommended 5 V supplies. The decibel gain is linear in dB, accurately calibrated, and stable over temperature and supply. The gain is controlled at a high
FUNCTIONAL BLOCK DIAGRAM
VPOS 8 VNEG 6 GPOS 1 VG GNEG 2 GAINCONTROL INTERFACE 6.44k1
5
SCALING REFERENCE
PRECISION PASSIVE INPUT ATTENUATOR
FIXED-GAIN AMPLIFIER
7
VOUT
AD603
FDBK
6941 0dB VINP 3 R 2R COMM 4 R-2R LADDER NETWORK
00539-001
-6.02dB R
-12.04dB -18.06dB -24.08dB R 2R 2R R 2R R
-30.1dB R 2R
-36.12dB -42.14dB R 2R R 201
1NOMINAL
VALUES.
Figure 1. Rev. G
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c) 2005 Analog Devices, Inc. All rights reserved.
AD603 TABLE OF CONTENTS
Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 4 ESD Caution.................................................................................. 4 Pin Configuration and Function Descriptions............................. 5 Typical Performance Characteristics ............................................. 6 Theory of Operation ...................................................................... 11 Noise Performance ..................................................................... 11 The Gain Control Interface....................................................... 12 Programming the Fixed-Gain Amplifier Using Pin Strapping ................................................................... 12 Using the AD603 in Cascade ........................................................ 14 Sequential Mode (Optimal S/N Ratio).................................... 14 Parallel Mode (Simplest Gain Control Interface) .................. 16 Low Gain Ripple Mode (Minimum Gain Error) ................... 16 Applications..................................................................................... 18 A Low Noise AGC Amplifier .................................................... 18 Caution ........................................................................................ 19 Outline Dimensions ....................................................................... 20 Ordering Guide .......................................................................... 20
REVISION HISTORY
3/05--Rev. F to Rev. G Updated Format.................................................................. Universal Change to Features ............................................................................1 Changes to General Description .....................................................1 Change to Figure 1 ............................................................................1 Changes to Specifications .................................................................3 New Figure 4 and Renumbering Subsequent Figures...................6 Change to Figure 10 ..........................................................................7 Change to Figure 23 ..........................................................................9 Change to Figure 29 ........................................................................12 Updated Outline Dimensions ........................................................20 4/04--Rev. E to Rev. F Changes to Specifications .................................................................2 Changes to Ordering Guide .............................................................3 8/03--Rev. D to Rev E Updated Format.................................................................. Universal Changes to Specifications .................................................................2 Changes to TPCs 2, 3, 4.....................................................................4 Changes to Sequential Mode (Optimal S/N Ratio) section .........9 Change to Figure 8 ..........................................................................10 Updated Outline Dimensions ........................................................14
Rev. G | Page 2 of 20
AD603 SPECIFICATIONS
@ TA = 25C, VS = 5 V, -500 mV VG +500 mV, GNEG = 0 V, -10 dB to +30 dB gain range, RL = 500 , and CL = 5 pF, unless otherwise noted. Table 1.
Parameter INPUT CHARACTERISTICS Input Resistance Input Capacitance Input Noise Spectral Density1 Noise Figure 1 dB Compression Point Peak Input Voltage OUTPUT CHARACTERISTICS -3 dB Bandwidth Slew Rate Peak Output2 Output Impedance Output Short-Circuit Current Group Delay Change vs. Gain Group Delay Change vs. Frequency Differential Gain Differential Phase Total Harmonic Distortion Third Order Intercept ACCURACY Gain Accuracy, f = 100 kHz; Gain (dB) = (40 VG + 10) dB TMIN to TMAX Gain, f = 10.7 MHz Conditions Pin 3 to Pin 4 Input short-circuited f = 10 MHz, gain = max, RS = 10 f = 10 MHz, gain = max, RS = 10 Min 97 Typ 100 2 1.3 8.8 -11 1.4 90 275 3.0 2 50 2 2 0.2 0.2 -60 15 0.5 -9.0 +10.5 +30.3 +1 +1.5 -8.0 +11.5 +31.3 +20 +30 +20 +30 40.6 42 39.9 +2.0 Max 103 Unit pF nV/Hz dB dBm V MHz V/s V mA ns ns % Degree dBc dBm dB dB dB dB dB mV mV mV mV dB/V dB/V dB/V V nA nA M dB/s V mA mA
2
VOUT = 100 mV rms RL 500 RL 500 f 10 MHz f = 3 MHz; full gain range VG = 0 V; f = 1 MHz to 10 MHz
2.5
f = 10 MHz, VOUT = 1 V rms f = 40 MHz, gain = max, RS = 50 -500 mV VG +500 mV, VG = -0.5 V VG = 0.0 V VG = 0.5 V VG = 0 V -500 mV VG +500 mV -1 -1.5 -10.3 +9.5 +29.3 -20 -30 -20 -30 39.4 38 38.7 -1.2
Output Offset Voltage3 TMIN to TMAX Output Offset Variation vs. VG TMIN to TMAX GAIN CONTROL INTERFACE Gain Scaling Factor TMIN to TMAX GNEG, GPOS Voltage Range4 Input Bias Current Input Offset Current Differential Input Resistance Response Rate POWER SUPPLY Specified Operating Range Quiescent Current TMIN to TMAX
1
100 kHz 10.7 MHz
40 39.3 200 10 50 80
Pin 1 to Pin 2 Full 40 dB gain change 4.75
12.5
6.3 17 20
Typical open or short-circuited input; noise is lower when system is set to maximum gain and input is short-circuited. This figure includes the effects of both voltage and current noise sources. 2 Using resistive loads of 500 or greater, or with the addition of a 1 k pull-down resistor when driving lower loads. 3 The dc gain of the main amplifier in the AD603 is x35.7; thus, an input offset of 100 V becomes a 3.57 mV output offset. 4 GNEG and GPOS, gain control, and voltage range are guaranteed to be within the range of -VS + 4.2 V to +VS - 3.4 V over the full temperature range of -40C to +85C.
Rev. G | Page 3 of 20
AD603 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Voltage VS Internal Voltage VINP (Pin 3) GPOS, GNEG (Pins 1, 2) Internal Power Dissipation1 Operating Temperature Range AD603A AD603S Storage Temperature Range Lead Temperature Range (Soldering 60 sec)
1
Rating 7.5 V 2 V Continuous VS for 10 ms VS 400 mW -40C to +85C -55C to +125C -65C to +150C 300C
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
Thermal Characteristics: 8-Lead SOIC Package: JA = 155C/W, JC = 33C/W, 8-Lead CERDIP Package: JA = 140C/W, JC = 15C/W.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. G | Page 4 of 20
AD603 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
GPOS 1 GNEG 2 VINP 3
8
VPOS VOUT
00539-002
GPOS 1 GNEG 2 VINP 3
8
AD603
7
Figure 2. 8-Lead Plastic SOIC (R) Package
Figure 3. 8-Lead Ceramic CERDIP (Q) Package
Table 3. Pin Function Descriptions
Pin No. 1 2 3 4 5 6 7 8 Mnemonic GPOS GNEG VINP COMM FDBK VNEG VOUT VPOS Description Gain Control Input High (Positive Voltage Increases Gain). Gain Control Input Low (Negative Voltage Increases Gain). Amplifier Input. Amplifier Ground. Connection to Feedback Network. Negative Supply Input. Amplifier Output. Positive Supply Input.
Rev. G | Page 5 of 20
00539-003
6 VNEG TOP VIEW COMM 4 (Not to Scale) 5 FDBK
VOUT TOP VIEW 6 VNEG (Not to Scale) 5 FDBK COMM 4
7
AD603
VPOS
AD603 TYPICAL PERFORMANCE CHARACTERISTICS
@ TA = 25C, VS = 5 V, -500 mV VG +500 mV, GNEG = 0 V, -10 dB to +30 dB gain range, RL = 500 , and CL = 5 pF, unless otherwise noted.
40
4 3 225 180 135 90 45 0 PHASE -2 -3 -4 -5
00539-004
30
2 1
20
0
GAIN (dB)
10.7MHz
-1
10
100kHz
-45 -90 -135 -180
0
-0.4
-0.2
0 VG (V)
0.2
0.4
0.6
1M
10M FREQUENCY (Hz)
100M
Figure 4. Gain vs. VG at 100 kHz and 10.7 mHz
2.5 2.0 1.5 45MHz 4 3 2 1 70MHz 10.7MHz 0.5 0 455kHz -0.5 70MHz -1.0
00539-005
Figure 7. Frequency and Phase Response vs. Gain (Gain = +10 dB, PIN = -30 dBm)
225 180 135 90 45 0 PHASE -2 -3 -4 -5 -6 100k -45 -90 -135 -180
00539-008
GAIN ERROR (dB)
1.0
0
GAIN (dB)
-1
-1.5 -0.5
-225 1M 10M FREQUENCY (Hz) 100M
-0.4
-0.3
-0.2 -0.1 0 0.1 0.2 GAIN VOLTAGE (Volts)
0.3
0.4
0.5
Figure 5. Gain Error vs. Gain Control Voltage at 455 kHz, 10.7 MHz, 45 MHz, 70 MHz
4 3 2 1 0
GAIN (dB)
Figure 8. Frequency and Phase Response vs. Gain (Gain = +30 dB, PIN = -30 dBm)
225 180 7.40 135 7.60
GAIN
45 0
GROUP DELAY (ns)
90
PHASE (Degrees)
7.20
-1 PHASE -2 -3 -4 -5 -6 100k
7.00
-45 -90 -135 -180
6.80
6.60
00539-006
-225 1M 10M FREQUENCY (Hz) 100M
-0.4
-0.2 0 0.2 GAIN CONTROL VOLTAGE (V)
0.4
0.6
Figure 6. Frequency and Phase Response vs. Gain (Gain = -10 dB, PIN = -30 dBm)
Figure 9. Group Delay vs. Gain Control Voltage
Rev. G | Page 6 of 20
00539-009
6.40 -0.6
PHASE (Degrees)
GAIN
00539-007
-10 -0.6
-6 100k
-225
PHASE (Degrees)
GAIN
GAIN (dB)
AD603
-1.0 -1.2
NEGATIVE OUTPUT VOLTAGE (V)
00539-010
-1.4 -1.6 -1.8 -2.0 -2.2 -2.4 -2.6 -2.8 -3.0 -3.2 0 50 100 200 500 1000 LOAD RESISTANCE () 2000
00539-013
+5V HP3326A DUALCHANNEL SYNTHESIZER 100
4
0.1F
8 3 5
AD603
2
1
7
10x PROBE 511
HP3585A SPECTRUM ANALYZER
6
0.1F -5V DATEL DVC 8500
-3.4
Figure 10. Third Order Intermodulation Distortion Test Setup
Figure 13. Typical Output Voltage Swing vs. Load Resistance (Negative Output Swing Limits First)
102
10dB/DIV
INPUT IMPEDANCE ()
00539-011
100
98
96
94
00539-014 00539-015
100k
1M
Figure 11. Third Order Intermodulation Distortion at 455 kHz (10x Probe Used to HP3585A Spectrum Analyzer, Gain = 0 dB, PIN = 0 dBm)
10M FREQUENCY (Hz)
100M
Figure 14. Input Impedance vs. Frequency (Gain = -10 dB)
10dB/DIV
102
INPUT IMPEDANCE ()
00539-012
100
98
96
94
100k
1M
Figure 12. Third Order Intermodulation Distortion at 10.7 MHz (10x Probe Used to HP3585A Spectrum Analyzer, Gain = 0 dB, PIN = 0 dBm)
10M FREQUENCY (Hz)
100M
Figure 15. Input Impedance vs. Frequency (Gain = +10 dB)
Rev. G | Page 7 of 20
AD603
3V
102
INPUT IMPEDANCE ()
100
INPUT GND 100MV/DIV
98
1V OUTPUT GND 1V/DIV
96
94
00539-016
100k
1M
10M FREQUENCY (Hz)
100M
50ns
451ns
Figure 16. Input Impedance vs. Frequency (Gain = +30 dB)
Figure 19. Output Stage Overload Recovery Time (Input Is 500 ns Period, 50% Duty Cycle Square Wave, Output is Captured Using Tektronix 11402 Digitizing Oscilloscope)
3.5V
1V
100 90
INPUT 500mV/DIV GND
500mV OUTPUT 500mV/DIV
10 0%
00539-017
GND
1V
200ns
50ns
456ns
Figure 17. Gain Control Channel Response Time
Figure 20. Transient Response, G = 0 dB (Input is 500 ns Period, 50% Duty Cycle Square Wave, Output is Captured Using Tektronix 11402 Digitizing Oscilloscope)
3.5V
4.5V
INPUT GND 1V/DIV
INPUT GND 100mV/DIV
500mV
500mV OUTPUT GND 500mV/DIV
OUTPUT GND 500mV/DIV
00539-018
50ns
451ns
50ns
456ns
Figure 18. Input Stage Overload Recovery Time (Input is 500 ns Period, 50% Duty Cycle Square Wave, Output is Captured Using Tektronix 11402 Digitizing Oscilloscope)
Figure 21. Transient Response, G = +20 dB (Input is 500 ns Period, 50% Duty Cycle Square Wave, Output is Captured Using Tektronix 11402 Digitizing Oscilloscope)
Rev. G | Page 8 of 20
00539-021
-500mV -49ns
-1.5V -44ns
00539-020
-1.5V -44ns
00539-019
-2V -49ns
AD603
21
0 -10
19 17
10MHz
TA = 25C RS = 50 TEST SETUP FIGURE 23
-20
NOISE FIGURE (dB)
PSRR (dB)
-30 -40 -50 -60
15 13 11 9 7
20MHz
00539-022
100k
1M
10M FREQUENCY (Hz)
100M
31
32
33
34
35 36 GAIN (dB)
37
38
39
40
Figure 22. PSRR vs. Frequency (Worst Case is Negative Supply PSRR, Shown Here)
0
Figure 25. Noise Figure in 0 dB/40 dB Mode
TA = 25C TEST SETUP FIGURE 23
+5V HP3326A DUALCHANNEL SYNTHESIZER 100
4
0.1F
-5
8 3 5
INPUT LEVEL (dBm)
-10
AD603
2
1
50
7
HP3585A SPECTRUM ANALYZER
-15
6
0.1F
00539-023
-5V DATEL DVC 8500
-20
30 50 INPUT FREQUENCY (MHz)
70
Figure 23. Test Setup Used for: Noise Figure, Third Order Intercept and 1 dB Compression Point Measurements
23 21 19 70MHz TA = 25C RS = 50V TEST SETUP FIGURE 23
Figure 26. 1 dB Compression Point, -10 dB/+30 dB Mode, Gain = +30 dB
20
TA = 25C TEST SETUP FIGURE 23 30MHz
18
NOISE FIGURE (dB)
17 50MHz 15 13 30MHz 11 9 7
00539-024
OUTPUT LEVEL (dBm)
16 40MHz 14
10MHz
12 70MHz 10
21
22
23
24
25 26 GAIN (dB)
27
28
29
30
-10 INPUT LEVEL (dBm)
0
Figure 24. Noise Figure in -10 dB/+30 dB Mode
Figure 27. Third Order Intercept -10 dB/+30 dB Mode, Gain = +10 dB
Rev. G | Page 9 of 20
00539-027
5 20
0 -20
00539-026
-25 10
00539-025
5 30
AD603
20 TA = 25C RS = 50 RIN = 50 RL = 100 TEST SETUP FIGURE 23
18 30MHz
OUTPUT LEVEL (dBm)
16 40MHz 14
12 70MHz 10
-30 INPUT LEVEL (dBm)
-20
Figure 28. Third Order Intercept -10 dB/+30 dB Mode, Gain = +30 dB
00539-028
8 -40
Rev. G | Page 10 of 20
AD603 THEORY OF OPERATION
The AD603 comprises a fixed-gain amplifier, preceded by a broadband passive attenuator of 0 dB to 42.14 dB, having a gain control scaling factor of 40 dB per volt. The fixed gain is lasertrimmed in two ranges, to either 31.07 dB (x35.8) or 50 dB (x358), or may be set to any range in between using one external resistor between Pin 5 and Pin 7. Somewhat higher gain can be obtained by connecting the resistor from Pin 5 to common, but the increase in output offset voltage limits the maximum gain to about 60 dB. For any given range, the bandwidth is independent of the voltage-controlled gain. This system provides an underrange and overrange of 1.07 dB in all cases; for example, the overall gain is -11.07 dB to +31.07 dB in the maximum bandwidth mode (Pin 5 and Pin 7 strapped). This X-AMP structure has many advantages over former methods of gain control based on nonlinear elements. Most importantly, the fixed-gain amplifier can use negative feedback to increase its accuracy. Since large inputs are first attenuated, the amplifier input is always small. For example, to deliver a 1 V output in the -1 dB/+41 dB mode (that is, using a fixed amplifier gain of 41.07 dB) its input is only 8.84 mV; thus the distortion can be very low. Equally important, the small-signal gain and phase response, and thus the pulse response, are essentially independent of gain. Figure 29 is a simplified schematic. The input attenuator is a seven-section R-2R ladder network, using untrimmed resistors of nominally R = 62.5 , which results in a characteristic resistance of 125 20%. A shunt resistor is included at the input and laser trimmed to establish a more exact input resistance of 100 3%, which ensures accurate operation (gain and HP corner frequency) when used in conjunction with external resistors or capacitors. The nominal maximum signal at input VINP is 1 V rms (1.4 V peak) when using the recommended 5 V supplies, although operation to 2 V peak is permissible with some increase in HF distortion and feedthrough. Pin 4 (COMM) must be connected directly to the input ground; significant impedance in this connection will reduce the gain accuracy. The signal applied at the input of the ladder network is attenuated by 6.02 dB by each section; thus, the attenuation to each of the taps is progressively 0 dB, 6.02 dB, 12.04 dB, 18.06 dB, 24.08 dB, 30.1 dB, 36.12 dB, and 42.14 dB. A unique circuit technique is employed to interpolate between these tap points, indicated by the slider in Figure 29, thus providing continuous attenuation from 0 dB to 42.14 dB. It will help in understanding the AD603 to think in terms of a mechanical means for moving this slider from left to right; in fact, its position is controlled by the voltage between Pin 1 and Pin 2. The details of the gain control interface are discussed later. The gain is at all times very exactly determined, and a linear-indB relationship is automatically guaranteed by the exponential nature of the attenuation in the ladder network (the X-AMP principle). In practice, the gain deviates slightly from the ideal law, by about 0.2 dB peak (see, for example, Figure 5).
NOISE PERFORMANCE
An important advantage of the X-AMP is its superior noise performance. The nominal resistance seen at inner tap points is 41.7 (one third of 125 ), which exhibits a Johnson noise spectral density (NSD) of 0.83 nV/Hz (that is, 4kTR) at 27C, which is a large fraction of the total input noise. The first stage of the amplifier contributes a further 1 nV/Hz, for a total input noise of 1.3 nV/Hz. It will be apparent that it is essential to use a low resistance in the ladder network to achieve the very low specified noise level. The signal's source impedance forms a voltage divider with the AD603's 100 input resistance. In some applications, the resulting attenuation may be unacceptable, requiring the use of an external buffer or preamplifier to match a high impedance source to the low impedance AD603. The noise at maximum gain (that is, at the 0 dB tap) depends on whether the input is short-circuited or open-circuited: when shorted, the minimum NSD of slightly over 1 nV/Hz is achieved; when open, the resistance of 100 looking into the first tap generates 1.29 nV/Hz, so the noise increases to a total of 1.63 nV/Hz. (This last calculation would be important if the AD603 were preceded by, for example, a 900 resistor to allow operation from inputs up to 10 V rms.) As the selected tap moves away from the input, the dependence of the noise on source impedance quickly diminishes. Apart from the small variations just discussed, the signal-tonoise (S/N) ratio at the output is essentially independent of the attenuator setting. For example, on the -11 dB/+31 dB range, the fixed gain of x35.8 raises the output NSD to 46.5 nV/Hz. Thus, for the maximum undistorted output of 1 V rms and a 1 MHz bandwidth, the output S/N ratio would be 86.6 dB, that is, 20 log (1 V/46.5 V).
Rev. G | Page 11 of 20
AD603
VPOS 8 VNEG 6 GPOS 1 VG GNEG 2 GAINCONTROL INTERFACE 6.44k1
5
SCALING REFERENCE
PRECISION PASSIVE INPUT ATTENUATOR
FIXED-GAIN AMPLIFIER
7
VOUT
AD603
FDBK
6941 0dB VINP 3 R 2R COMM 4 R-2R LADDER NETWORK
00539-029
-6.02dB R
-12.04dB -18.06dB -24.08dB R 2R 2R R 2R R
-30.1dB R 2R
-36.12dB -42.14dB R 2R R 201
1NOMINAL
VALUES.
Figure 29. Simplified Block Diagram
THE GAIN CONTROL INTERFACE
The attenuation is controlled through a differential, high impedance (50 M) input, with a scaling factor which is lasertrimmed to 40 dB per volt, that is, 25 mV/dB. An internal band gap reference ensures stability of the scaling with respect to supply and temperature variations. When the differential input voltage VG = 0 V, the attenuator slider is centered, providing an attenuation of 21.07 dB. For the maximum bandwidth range, this results in an overall gain of 10 dB (= -21.07 dB + 31.07 dB). When the control input is -500 mV, the gain is lowered by 20 dB (= 0.500 V x 40 dB/V) to -10 dB; when set to +500 mV, the gain is increased by 20 dB to 30 dB. When this interface is overdriven in either direction, the gain approaches either -11.07 dB (= - 42.14 dB + 31.07 dB) or 31.07 dB (= 0 + 31.07 dB), respectively. The only constraint on the gain control voltage is that it be kept within the commonmode range (-1.2 V to +2.0 V assuming +5 V supplies) of the gain control interface. The basic gain of the AD603 can thus be calculated using the following simple expression: Gain (dB) = 40 VG +10 (1)
For example, if the gain is to be controlled by a DAC providing a positive only ground-referenced output, the Gain Control Low (GNEG) pin should be biased to a fixed offset of 500 mV to set the gain to -10 dB when Gain Control High (GPOS) is at zero, and to 30 dB when at 1.00 V. It is a simple matter to include a voltage divider to achieve other scaling factors. When using an 8-bit DAC having an FS output of 2.55 V (10 mV/bit), a divider ratio of 2 (generating 5 mV/bit) would result in a gain-setting resolution of 0.2 dB/bit. The use of such offsets is valuable when two AD603s are cascaded, when various options exist for optimizing the S/N profile, as will be shown later.
PROGRAMMING THE FIXED-GAIN AMPLIFIER USING PIN STRAPPING
Access to the feedback network is provided at Pin 5 (FDBK). The user may program the gain of the AD603's output amplifier using this pin, as shown in Figure 30, Figure 31, and Figure 32. There are three modes: in the default mode, FDBK is unconnected, providing the range +9 dB/+51 dB; when VOUT and FDBK are shorted, the gain is lowered to -11 dB/+31 dB; and when an external resistor is placed between VOUT and FDBK any intermediate gain can be achieved, for example, -1 dB/+41 dB. Figure 33 shows the nominal maximum gain vs. external resistor for this mode.
where VG is in volts. When Pin 5 and Pin 7 are strapped (see next section), the gain becomes Gain (dB) = 40 VG + 20 for 0 to +40 dB and Gain (dB) = 40 VG + 30 for +10 to +50 dB (2)
VC1
1
GPOS
VPOS 8
VPOS
AD603
VC2
2
GNEG
VOUT 7
VOUT
The high impedance gain control input ensures minimal loading when driving many amplifiers in multiple channel or cascaded applications. The differential capability provides flexibility in choosing the appropriate signal levels and polarities for various control schemes.
Rev. G | Page 12 of 20
VIN
3
VINP
VNEG 6
VNEG
00539-030
4
COMM
FDBK 5
Figure 30. -10 dB to +30 dB; 90 MHz Bandwidth
AD603
VC1
1
GPOS
VPOS 8
VPOS
AD603
VC2
2
GNEG
VOUT 7
VOUT
VIN
3
VINP
VNEG 6
VNEG
2.15k
00539-031
Optionally, when a resistor is placed from FDBK to COMM, higher gains can be achieved. This fourth mode is of limited value because of the low bandwidth and the elevated output offsets; it is thus not included in Figure 30, Figure 31, or Figure 32. The gain of this amplifier in the first two modes is set by the ratio of on-chip laser-trimmed resistors. While the ratio of these resistors is very accurate, the absolute value of these resistors can vary by as much as 20%. Thus, when an external resistor is connected in parallel with the nominal 6.44 k 20% internal resistor, the overall gain accuracy is somewhat poorer. The worst-case error occurs at about 2 k (see Figure 34).
4
COMM
FDBK 5
5.6pF
Figure 31. 0 dB to 40 dB; 30 MHz Bandwidth
VC1
1
GPOS
VPOS 8
VPOS
AD603
VC2
2
GNEG
VOUT 7
VOUT
1.2 1.0
00539-032
VIN
3
VINP
VNEG 6
VNEG
-1:VdB (OUT) - (-1):VdB (OREF)
4
COMM
FDBK 5
0.8 0.6 0.4
DECIBELS
18pF
Figure 32. 10 dB to 50 db; 9 MHz to Set Gain
0.2 0 -0.2 -0.4 -0.6 -0.8 100 1k REXT () 10k 100k 1M
00539-034
52 50 48 46 44
DECIBELS
-1:VdB (OUT)
42 40 38 36 34 32 100
VdB (OUT) -2:VdB (OUT)
-1.0 10
VdB (OUT) - VdB (OREF)
Figure 34. Worst-Case Gain Error, Assuming Internal Resistors have a Maximum Tolerance of -20% (Top Curve) or =20% (Bottom Curve)
1k REXT ()
10k
100k
1M
Figure 33. Gain vs. REXT, Showing Worst-Case Limits Assuming Internal Resistors have a Maximum Tolerance of 20%
00539-033
30 10
While the gain bandwidth product of the fixed-gain amplifier is about 4 GHz, the actual bandwidth is not exactly related to the maximum gain. This is because there is a slight enhancing of the ac response magnitude on the maximum bandwidth range, due to higher order poles in the open-loop gain function; this mild peaking is not present on the higher gain ranges. Figure 30, Figure 31, and Figure 32 show how an optional capacitor may be added to extend the frequency response in high gain modes.
Rev. G | Page 13 of 20
AD603 USING THE AD603 IN CASCADE
Two or more AD603s can be connected in series to achieve higher gain. Invariably, ac coupling must be used to prevent the dc offset voltage at the output of each amplifier from overloading the following amplifier at maximum gain. The required high-pass coupling network will usually be just a capacitor, chosen to set the desired corner frequency in conjunction with the well-defined 100 input resistance of the following amplifier. For two AD603s, the total gain control range becomes 84 dB (2 x 42.14 dB); the overall -3 dB bandwidth of cascaded stages will be somewhat reduced. Depending on the pin strapping, the gain and bandwidth for two cascaded amplifiers can range from -22 dB to +62 dB (with a bandwidth of about 70 MHz) to +22 dB to +102 dB (with a bandwidth of about 6 MHz). There are several ways of connecting the gain control inputs in cascaded operation. The choice depends on whether it is important to achieve the highest possible instantaneous signalto-noise ratio (ISNR), or, alternatively, to minimize the ripple in the gain error. The following examples feature the AD603 programmed for maximum bandwidth; the explanations apply to other gain/bandwidth combinations with appropriate changes to the arrangements for setting the maximum gain. be provided by resistive dividers operating from a common voltage reference.
90 85 80
S/N RATIO (dB)
75 70 65 60 55
00539-035
50 -0.2
0.2
0.6
1.0 VC (V)
1.4
1.8
2.2
Figure 35. SNR vs. Control Voltage-Sequential Control (1 MHz Bandwidth)
SEQUENTIAL MODE (OPTIMAL S/N RATIO)
In the sequential mode of operation, the ISNR is maintained at its highest level for as much of the gain control range as possible. Figure 35 shows the SNR over a gain range of -22 dB to +62 dB, assuming an output of 1 V rms and a 1 MHz bandwidth; Figure 36, Figure 37, and Figure 38 show the general connections to accomplish this. Here, both the positive gain control inputs (GPOS) are driven in parallel by a positive-only, ground-referenced source with a range of 0 V to +2 V, while the negative gain control inputs (GNEG) are biased by stable voltages to provide the needed gain offsets. These voltages may
A1
-40.00dB
The gains are offset (Figure 39) such that A2's gain is increased only after A1's gain has reached its maximum value. Note that for a differential input of -600 mV or less, the gain of a single amplifier (A1 or A2) will be at its minimum value of -11.07 dB; for a differential input of +600 mV or more, the gain will be at its maximum value of 31.07 dB. Control inputs beyond these limits will not affect the gain and can be tolerated without damage or foldover in the response. This is an important aspect of the AD603's gain control response. (See the Specifications section for more details on the allowable voltage range.) The gain is now Gain (dB) = 40 VG + GO (3)
where VG is the applied control voltage and GO is determined by the gain range chosen. In the explanatory notes that follow, it is assumed the maximum bandwidth connections are used, for which GO is -20 dB.
A2 -51.07dB
INPUT 0dB
-42.14dB GPOS VG1 GNEG
-8.93dB 31.07dB
-42.14dB GPOS VG2 GNEG
31.07dB
VC = 0V
VO1 = 0.473V
VO2 = 1.526V
Figure 36. AD603 Gain Control Input Calculations for Sequential Control Operation VC = 0 V
0dB -11.07dB
INPUT 0dB
0dB GPOS VG1 GNEG
31.07dB 31.07dB
-42.14dB GPOS VG2 GNEG
31.07dB
VC = 1.0V
VO1 = 0.473V
VO2 = 1.526V
Figure 37. AD603 Gain Control Calculations for Sequential Control Operation VC = 1.0 V
Rev. G | Page 14 of 20
00539-037
OUTPUT 20dB
00539-036
OUTPUT -20dB
AD603
0dB -28.93dB
INPUT 0dB
0dB GPOS VG1 GNEG
31.07dB 31.07dB
-2.14dB GPOS VG2 GNEG
31.07dB
VC = 2.0V
VO1 = 0.473V
VO2 = 1.526V
Figure 38. AD603 Gain Control Input Calculations for Sequential Operation VC = 2.0 V
+31.07dB +31.07dB +10dB -8.93dB +28.96dB
70 60 COMBINED 50
OVERALL GAIN (dB)
A1
1 1
A2
-11.07dB -11.07dB 0.473 1.526 0.5 0 1.0 20 1.50 40 2.0 60 VC (V) 62.14
00539-039
40 A1 30 20 10 0 A2 -10 -20
00539-040 00539-041
GAIN (dB) -22.14
0 -20
1GAIN
OFFSET OF 1.07dB, OR 26.75mV.
Figure 39. Explanation of Offset Calibration for Sequential Control
With reference to Figure 36, Figure 37, and Figure 38, note that VG1 refers to the differential gain control input to A1, and VG2 refers to the differential gain control input to A2. When VG is 0, VG1 = -473 mV and thus the gain of A1 is -8.93 dB (recall that the gain of each individual amplifier in the maximum bandwidth mode is -10 dB for VG = -500 mV and 10 dB for VG = 0 V); meanwhile, VG2 = -1.908 V so the gain of A2 is pinned at -11.07 dB. The overall gain is thus -20 dB. See Figure 36. When VG = +1.00 V, VG1 = 1.00 V - 0.473 V = +0.526 V, which sets the gain of A1 to at nearly its maximum value of 31.07 dB, while VG2 = 1.00 V - 1.526 V = 0.526 V, which sets A2's gain at nearly its minimum value of -11.07 dB. Close analysis shows that the degree to which neither AD603 is completely pushed to its maximum nor minimum gain exactly cancels in the overall gain, which is now +20 dB. See Figure 37. When VG = 2.0 V, the gain of A1 is pinned at 31.07 dB and that of A2 is near its maximum value of 28.93 dB, resulting in an overall gain of 60 dB (see Figure 38). This mode of operation is further clarified by Figure 40, which is a plot of the separate gains of A1 and A2 and the overall gain vs. the control voltage. Figure 41 is a plot of the SNR of the cascaded amplifiers vs. the control voltage. Figure 42 is a plot of the gain error of the cascaded stages vs. the control voltages.
S/N RATIO (dB)
-30 -0.2
0.2
0.6
1.0 VC
00539-038
OUTPUT 60dB
1.4
1.8
2.0
Figure 40. Plot of Separate and Overall Gains in Sequential Control
90 80 70 60 50 40 30 20 10 -0.2
0.2
0.6
1.0 VC
1.4
1.8
2.0
Figure 41. SNR for Cascaded Stages--Sequential Control
Rev. G | Page 15 of 20
AD603
2.0 1.5 1.0
GAIN ERROR (dB) IS/N RATIO (dB)
00539-042
90 85 80 75 70 65 60 55
00539-044 00539-045
0.5 0 -0.5 -1.0 -1.5 -2.0 -0.2
0
0.2
0.4
0.6
0.8
1.0 VC
1.2
1.4
1.6
1.8
2.0
2.2
50 -0.2
0
0.2
0.4 VC
0.6
0.8
1.0
1.2
Figure 42. Gain Error for Cascaded Stages-Sequential Control
Figure 44. ISNR for Cascaded Stages--Parallel Control
PARALLEL MODE (SIMPLEST GAIN CONTROL INTERFACE)
In this mode, the gain control of voltage is applied to both inputs in parallel--the GPOS pins of both A1 and A2 are connected to the control voltage and the GNEW inputs are grounded. The gain scaling is then doubled to 80 dB/V, requiring only a 1.00 V change for an 80 dB change of gain: Gain = (dB) = 80 VG + GO (4)
LOW GAIN RIPPLE MODE (MINIMUM GAIN ERROR)
As can be seen in Figure 42 and Figure 43, the error in the gain is periodic, that is, it shows a small ripple. (Note that there is also a variation in the output offset voltage, which is due to the gain interpolation, but this is not exact in amplitude.) By offsetting the gains of A1 and A2 by half the period of the ripple, that is, by 3 dB, the residual gain errors of the two amplifiers can be made to cancel. Figure 45 shows much lower gain ripple when configured in this manner. Figure 46 plots the ISNR as a function of gain; it is very similar to that in the parallel mode.
3.0 2.5 2.0
where, as before, GO depends on the range selected; for example, in the maximum bandwidth mode, GO is +20 dB. Alternatively, the GNEG pins may be connected to an offset voltage of 0.500 V, in which case GO is -20 dB. The amplitude of the gain ripple in this case is also doubled, as shown in Figure 43, while the instantaneous signal-to-noise ratio at the output of A2 now decreases linearly as the gain increases, as shown in Figure 44.
2.0 1.5 1.0
GAIN ERROR (dB)
1.5
GAIN ERROR (dB)
1.0 0.5 0 -0.5 -1.0 -1.5 -2.0 -2.5
0.5 0 -0.5 -1.0 -1.5
00539-043
-3.0 -0.1
0
0.1
0.2
0.3
0.4
0.5 VC
0.6
0.7
0.8
0.9
1.0
1.1
Figure 45. Gain Error for Cascaded Stages--Low Ripple Mode
-2.0 -0.2
0
0.2
0.4
0.6
0.8
1.0 VC
1.2
1.4
1.6
1.8
2.0
2.2
Figure 43. Gain Error for Cascaded Stages--Parallel Control
Rev. G | Page 16 of 20
AD603
90 85 80
IS/N RATIO (dB)
75 70 65 60 55
00539-046
50 -0.2
0
0.2
0.4 VC
0.6
0.8
1.0
1.2
Figure 46. ISNR vs. Control Voltage--Low Ripple Mode
Rev. G | Page 17 of 20
AD603 APPLICATIONS
A LOW NOISE AGC AMPLIFIER
Figure 47 shows the ease with which the AD603 can be connected as an AGC amplifier. The circuit illustrates many of the points previously discussed: It uses few parts, has linear-indB gain, operates from a single supply, uses two cascaded amplifiers in sequential gain mode for maximum S/N ratio, and an external resistor programs each amplifier's gain. It also uses a simple temperature-compensated detector. The circuit operates from a single 10 V supply. Resistors R1, R2, R3, and R4 bias the common pins of A1 and A2 at 5 V. This pin is a low impedance point and must have a low impedance path to ground, provided here by the 100 F tantalum capacitors and the 0.1 F ceramic capacitors. The cascaded amplifiers operate in sequential gain. Here, the offset voltage between the Pin 2 (GNEG) of A1 and A2 is 1.05 V (42.14 dB x 25 mV/dB), provided by a voltage divider consisting of resistors R5, R6, and R7. Using standard values, the offset is not exact, but it is not critical for this application. The gain of both A1 and A2 is programmed by resistors R13 and R14, respectively, to be about 42 dB; thus the maximum gain of the circuit is twice that, or 84 dB. The gain control range can be shifted up by as much as 20 dB by appropriate choices of R13 and R14. The circuit operates as follows. A1 and A2 are cascaded. Capacitor C1 and the 100 of resistance at the input of A1 form a time constant of 10 s. C2 blocks the small dc offset voltage at the output of A1 (which might otherwise saturate A2 at its maximum gain) and introduces a high-pass corner at about 16 kHz, eliminating low frequency noise. A half-wave detector is used, based on Q1 and R8. The current into capacitor CAV is just the difference between the collector current of Q2 (biased to be 300 A at 300 K, 27C) and the collector current of Q1, which increases with the amplitude of the output signal. The automatic gain control voltage, VAGC, is the time integral of this error current. In order for VAGC (and thus the gain) to remain insensitive to short-term amplitude fluctuations in the output signal, the rectified current in Q1 must, on average, exactly balance the current in Q2. If the output of A2 is too small to do this, VAGC will increase, causing the gain to increase, until Q1 conducts sufficiently. Consider the case where R8 is zero and the output voltage VOUT is a square wave at, say, 455 kHz, which is well above the corner frequency of the control loop.
10V
C7 0.1F C1 0.1F J1 R T1 100 10V R1 2.49k C 32 100F + C4 0.1F
4 3
THIS CAPACITOR SETS AGC TIME CONSTANT 10V R1 3 2.49k
6
R9 1.54k Q2 2N3906
R1 0 1.24k
C11 0.1F
VAGC C8 0.1F C2 0.1F
7 2 3
10V R1 4 2.49k
6
8
CAV 0.1F
Q1 2N3904 R8 806
R1 1 3.83k 5V R1 2 4.99k C9 0.1F J2 C10 0.1F
A1 AD603
1
5
8
10V R3 2.49k C 52 100F + C6 0.1F
4
A2 AD603
1
5
7 2
R2 2.49k
R4 2.49k AGC LINE
R5 5.49k 5.5V
1V OFFSET FOR SEQUENTIAL GAIN R6 1.05k 6.5V
R7 3.48k 10V
1 RT 2C3
PR OVI D ES A 5 0 IN PU T IMPED A N C E. A N D C 5 A R E TA N TA LU M.
Figure 47. A Low Noise AGC Amplifier
Rev. G | Page 18 of 20
00539-047
AD603
During the time VOUT is negative with respect to the base voltage of Q1, Q1 conducts; when VOUT is positive, it is cut off. Since the average collector current of Q1 is forced to be 300 A, and the square wave has a duty cycle of 1:1, Q1's collector current when conducting must be 600 A. With R8 omitted, the peak amplitude of VOUT is forced to be just the VBE of Q1 at 600 A, typically about 700 mV, or 2 VBE peak-to-peak. This voltage, the amplitude at which the output stabilizes, has a strong negative temperature coefficient (TC), typically -1.7 mV/C. Although this may not be troublesome in some applications, the correct value of R8 will render the output stable with temperature. To understand this, note that the current in Q2 is made to be proportional to absolute temperature (PTAT). For the moment, continue to assume that the signal is a square wave. When Q1 is conducting, VOUT is now the sum of VBE and a voltage that is PTAT and that can be chosen to have an equal but opposite TC to that of the VBE. This is actually nothing more than an application of the band gap voltage reference principle. When R8 is chosen such that the sum of the voltage across it and the VBE of Q1 is close to the band gap voltage of about 1.2 V, VOUT will be stable over a wide range of temperatures, provided, of course, that Q1 and Q2 share the same thermal environment. Since the average emitter current is 600 A during each half cycle of the square wave, a resistor of 833 would add a PTAT voltage of 500 mV at 300 K, increasing by 1.66 mV/C. In practice, the optimum value will depend on the type of transistor used and, to a lesser extent, on the waveform for which the temperature stability is to be optimized; for the inexpensive 2N3904/2N3906 pair and sine wave signals, the recommended value is 806 . This resistor also serves to lower the peak current in Q1 when more typical signals (usually sinusoidal) are involved, and the 1.8 kHz LP filter it forms with CAV helps to minimize distortion due to ripple in VAGC. Note that the output amplitude under sine wave conditions will be higher than for a square wave, since the average value of the current for an ideal rectifier would be 0.637 times as large, causing the output amplitude to be 1.88 (= 1.2/0.637) V, or 1.33 V rms. In practice, the somewhat nonideal rectifier results in the sine wave output being regulated to about 1.4 V rms, or 3.6 V p-p. The bandwidth of the circuit exceeds 40 MHz. At 10.7 MHz, the AGC threshold is 100 V (-67 dBm) and its maximum gain is 83 dB (20 log 1.4 V/100 V). The circuit holds its output at 1.4 V rms for inputs as low as -67 dBm to +15 dBm (82 dB), where the input signal exceeds the AD603's maximum input rating. For a 30 dBm input at 10.7 MHz, the second harmonic is 34 dB down from the fundamental and the third harmonic is 35 dB down.
CAUTION
Careful component selection, circuit layout, power supply decoupling, and shielding are needed to minimize the AD603's susceptibility to interference from signals such as those from radio and TV stations. In bench evaluation, it is recommended to place all of the components into a shielded box and using feedthrough decoupling networks for the supply voltage. Circuit layout and construction are also critical, since stray capacitances and lead inductances can form resonant circuits and are a potential source of circuit peaking, oscillation, or both.
Rev. G | Page 19 of 20
AD603 OUTLINE DIMENSIONS
0.005 (0.13) MIN
8
0.055 (1.40) MAX
5
0.310 (7.87) 0.220 (5.59)
1 4
5.00 (0.1968) 4.80 (0.1890)
8 5 4
PIN 1
0.100 (2.54) BSC 0.405 (10.29) MAX 0.320 (8.13) 0.290 (7.37) 0.060 (1.52) 0.015 (0.38) 0.150 (3.81) MIN SEATING 0.070 (1.78) PLANE 0.030 (0.76) 15 0 0.015 (0.38) 0.008 (0.20)
4.00 (0.1574) 3.80 (0.1497) 1
6.20 (0.2440) 5.80 (0.2284)
0.200 (5.08) MAX 0.200 (5.08) 0.125 (3.18) 0.023 (0.58) 0.014 (0.36)
1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040)
1.75 (0.0688) 1.35 (0.0532)
0.50 (0.0196) x 45 0.25 (0.0099)
0.51 (0.0201) COPLANARITY SEATING 0.31 (0.0122) 0.10 PLANE
8 0.25 (0.0098) 0 1.27 (0.0500) 0.40 (0.0157) 0.17 (0.0067)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
COMPLIANT TO JEDEC STANDARDS MS-012-AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 48. 8-Lead Ceramic Dual In-Line Package [CERDIP] (Q-8) Dimensions shown in inches and (millimeters)
Figure 49. 8-Lead Standard Small Outline Package [SOIC-N] Narrow Body (R-8) Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Part Number AD603AR AD603AR-REEL AD603AR-REEL7 AD603ARZ1 AD603ARZ-REEL1 AD603ARZ-REEL71 AD603AQ AD603SQ/883B2 AD603-EB AD603ACHIPS
1 2
Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -55C to +125C
Package Description 8-Lead SOIC 8-Lead SOIC, 13" Reel 8-Lead SOIC, 7" Reel 8-Lead SOIC 8-Lead SOIC, 13" Reel 8-Lead SOIC, 7" Reel 8-Lead CERDIP 8-Lead CERDIP Evaluation Board DIE
Package Option R-8 R-8 R-8 R-8 R-8 R-8 Q-8 Q-8
Z = Pb-free part. Refer to AD603 Military data sheet. Also available as 5962-9457203MPA.
(c) 2005 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. C00539-0-3/05(G)
Rev. G | Page 20 of 20

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